Resampling an audio signal for low-delay encoding/decoding

ABSTRACT

A method and device for resampling an audio frequency signal in an audio frequency signal coding or decoding. The method includes the following acts for each signal block to be resampled: determining, by adaptive linear prediction, a number of future signal samples, this number being defined as a function of a chosen resampling delay; constructing a resampling support vector from at least samples of the current block and determined future signal samples; applying a resampling filter to the samples of the resampling support vector.

CROSS-REFERENCE TO RELATED APPLICATIONS

This Application is a Section 371 National Stage Application ofInternational Application No. PCT/FR2014/052430, filed Sep. 26, 2014,the content of which is incorporated herein by reference in itsentirety, and published as WO 2015/044609 on Apr. 2, 2015, not inEnglish.

FIELD OF THE DISCLOSURE

The present invention relates to the processing of an audio frequencysignal for the transmission or storage thereof. More particularly, theinvention relates to a change of sampling frequency upon a coding or adecoding of the audio frequency signal.

BACKGROUND OF THE DISCLOSURE

There are many techniques for compressing (with loss) an audio frequencysignal such as speech or music. The coding can be performed directly atthe sampling frequency of the input signal, as for example in the ITU-Trecommendations G.711 or G.729 in which the input signal is sampled at 8kHz and the coder and decoder operate at this same frequency.

However, some coding methods use a change of sampling frequency, forexample to reduce the complexity of the coding, adapt the codingaccording to the different frequency subbands to be coded, or convertthe input signal for it to correspond to a predefined internal samplingfrequency of the coder.

In the subband coding defined in the ITU-T recommendation G.722, theinput signal at 16 kHz is divided into 2 subbands (sampled at 8 kHz)which are coded separately by a coder of ADPCM (Adaptive DifferentialPulse Code Modulation) type. This division into two subbands is carriedout by a bank of quadratic mode mirror filters with Finite ImpulseResponse (FIR), of order 23, which theoretically results in ananalysis-synthesis delay (coder+decoder) of 23 samples at 16 ms; thisfilter bank is implemented with a polyphase realization. The divisioninto two subbands in G.722 makes it possible to allocate, in apredetermined manner, different bit rates to the two subbands as afunction of their a priori perceptual importance and also to reduce theoverall coding complexity by executing two coders of ADPCM type at alower frequency. However, it induces an algorithmic delay compared to adirect ADPCM coding.

Various methods for changing the sampling frequency, also calledresampling, of a digital signal are known, by using, for example and ina nonexhaustive manner, an FIR (Finite Impulse Response) filter, an IIR(Infinite Impulse Response) filter or a polynominal interpolation(including the splines). A review of the conventional resampling methodscan be found for example in the article by R. W. Schafer, L. R. Rabiner,A Digital Signal Processing Approach to Interpolation, Proceedings ofthe IEEE, vol. 61, no. 6, June 1973, pp. 692-702.

The advantage of the FIR filter (symmetrical) lies in its simplifiedimplementation and—subject to certain conditions—in the possibility ofensuring a linear phase. A linear phase filtering makes it possible topreserve the waveform of the input signal, but it can also beaccompanied by a temporal spreading (or ringing) that can createartifacts of pre-echo type on transients. This method results in a delay(which is dependent on the length of the impulse response), generally ofthe order of 1 to a few ms to ensure appropriate filteringcharacteristics (ripple in the bandwidth, rejection level sufficient toeliminate the aliasing or spectral images . . . ).

The alternative of resampling by IIR filter generally leads to anon-linear phase, unless the phase is compensated by an additionalall-pass filtering stage as described for example in the article by P.A. Regalia, S. K. Mitra, P. P. Vaidyanathan, The Digital All-PassFilter: A Versatile Signal Processing Building Block, Proceedings of theIEEE, vol. 76, no. 1, January 1988, with an exemplary realization in the“iirgrpdelay” routine of the MATLAB software; an IIR filter is generallyof a lower order but more complex to implement in fixed-point notation,the states (or memories) of the filter being able to reach high dynamicvalues for the recursive part, and this problem is amplified if a phasecompensation by all-pass filtering is used.

FIG. 1 illustrates an example of down-sampling by a ratio of 4/5 with anFIR filter with a length of 2*60+1=121 coefficients at 64 kHz to changefrom 16 kHz to 12.8 kHz. The x-axes represent the time (grounded to msto represent the signals clocked at different frequencies) and they-axes the amplitudes. The squares at the top represent the temporalpositions of the samples of the input signal at 16 kHz; it is assumedhere that these samples correspond to the end of a 20 ms frame. Thecontinuous vertical lines mark the corresponding sampling instants at 16kHz. At the bottom of the figure, the dotted vertical lines mark thecorresponding sampling instants at 12.8 kHz and the stars symbolize theoutput samples at 12.8 kHz. Also represented is the impulse response(symmetrical) of 121 coefficients of an FIR filter at 64 kHz, thisresponse is positioned to calculate the last sample of the current frameat 12.8 kHz (the position of the impulse response maximum is alignedwith this sample). The circles show the values used (corresponding tothe input sampling moment) in a polyphase representation; to obtain theoutput sample, these values are multiplied by the corresponding inputsample and these results are added together. It will be noted in thisfigure that 12 samples (up to the end of the input frame) at 12.8 kHzcannot be calculated exactly because the input samples after the end ofthe current frame (start of the next frame) are not yet known; thedown-sampling delay in the conditions of FIG. 1 is 12 samples, i.e.12/12.8=0.9375 ms.

There are techniques for reducing the delay introduced by the changes ofsampling frequency of FIR type.

In the 3GPP AMR-WB standard (also defined as the ITU-T recommendationG.722.2), the input signal sampled at 16 kHz is down-sampled at aninternal frequency of 12.8 kHz before applying a coding of CELP type;the signal decoded at 12.8 kHz is then resampled at 16 kHz and combinedwith a high-band signal.

The advantage of passing through an intermediate frequency of 12.8 kHzis that it makes it possible to reduce the complexity of the CELP codingand also to have a frame length that is a multiple of a power of 2,which facilitates the coding of certain CELP parameters. The method usedis a conventional resampling by a factor 4/5 by FIR filter (of 121coefficients at 64 kHz), with a polyphase realization to minimize thecomplexity.

In theory, this resampling on the coder and on the AMR-WB decoder shouldresult in a delay in a manner similar to the processing represented inFIG. 1. In the case of the AMR-WB codec, with an FIR filter of 121coefficients, the total delay should in theory be 2×60 samples at 64kHz, i.e. 2×15 samples at 16 kHz or 1.875 ms; in fact, a specifictechnique is implemented on the AMR-WB coder to eliminate (compensate)the associated delay in the coder part only and therefore divide theeffective delay by 2.

This compensation method is described in the 3GPP standard TS 26.190,Clause 5.1 and in the ITU-T recommendation G.722.2, Clause 5.1. Themethod for compensating the FIR filtering delay consists in adding, foreach new frame sampled at 16 kHz to be converted to 12.8 kHz, apredetermined number of zeros at the end of the current frame. Thesezeros are defined at the input sampling frequency and their numbercorresponds to the delay of the resampling FIR filter at this frequency(i.e. 15 zeros added at 16 kHz). The resampling is implemented per 20 msframe (320 samples). The resampling in the AMR-WB coder is thereforeequivalent to complementing the input frame of 320 samples on the left(toward the past) with 30 samples from the end of preceding frame(resampling memory) and on the right with 15 zeros to form a vector of30+320+15=365 samples, which is then resampled with a factor 4/5. TheFIR filter can thus be implemented with a zero phase, therefore withoutdelay, since a null future signal is added. In theory, the FIRresampling by a factor 4/5 is performed according to the followingsteps:

-   -   up-sampling by 4 (from 16 kHz to 64 kHz) by addition of 3        samples at 0 after each input sample    -   low-pass filtering of transfer function H_(decim)(z) of        symmetrical FIR type of order 120 at 64 kHz    -   down-sampling by 5 (from 64 kHz to 12.8 kHz) by keeping only one        sample out of five from the low-pass filtered signal.

In practice, this resampling is implemented in an equivalent manneraccording to an optimized polyphase realization without calculating theintermediate signal at 64 kHz and without concatenating the signal to beconverted with zeros (see the “decim54.c” file of the source code of theAMR-WB codec); the FIR filtering for each “phase” is equivalent to anFIR filter of order 24 at 12.8 kHz with a delay of 12 samples at 12.8kHz, i.e. 0.9375 ms.

The result of the FIR resampling of each 20 ms frame from 16 kHz to 12.8kHz is identical to a resampling formed on the “complete” input signal(i.e. not cut up into frames), except for the last 12 samples of eachresulting frame at 12.8 kHz which include an error due to the use of ablock of zeros as future signal instead of the “true” future signalwhich is available only on the next frame. In fact, the zeros introducedsimulate the case of a null input signal in the next frame.

This processing is illustrated at the end of a 20 ms frame in FIG. 2which represents the last input samples at 16 kHz by the squares at thetop; the vertical lines mark the corresponding sampling moments at 16kHz. At the bottom of the figure, the stars symbolize the output samplesat 12.8 kHz which can be obtained by conventional down-sampling with adelay of 12 samples. Then, the triangles at the bottom correspond to the12 samples at 12.8 kHz obtained by using at least one sample of nullvalue added at the end of the frame to be able to continue the filteringand reduce the delay. These samples are numbered from #1 to #12according to their position relative to the end of the output obtainedwith a conventional filtering. Also represented is the impulse responseof the filter at 64 kHz used in the position corresponding to the lastoutput sample at 12.8 kHz (the impulse response maximum is aligned withthis sample). The circles show the values used (corresponding to theinput sampling moment) in the polyphase representation; to obtain theoutput sample, these values are multiplied by the corresponding inputsample or by 0 for the values after the end of the frame and theseresults are added together. It can be seen here that, for this lastsample, almost half of the samples used from the impulse response aremultiplied by the added zeros, which therefore introduces a significantestimation error. It will also be understood that the error of the firstsamples generated after the conventional filtering (that is to say withonly the true input signal) is small (the weight of the impulse responseat its end is low) and the error becomes greater with increasingdistance from the conventional filtering (the weight of the impulseresponse then being greater). That will be able to be observed in theresults of FIG. 7.

The delay compensation method used in the AMR-WB codec, where zeros areadded at the end of each 20 ms block (or frame) to be resampled, makesit possible to eliminate the resampling delay on the coder, but it isnot satisfactory generally when the values generated at the end of thecurrent frame (with zeros added at the input) are coded directly and arenot replaced by the true values once the input signal of the next frameis known. In fact, these regular errors at the end of each framegenerate periodic discontinuities in the transition to the true outputsignal at the start of the next frame. These discontinuities are oftenaudible and a great nuisance. This is why the delay compensation isapplied only on the coder and only in the future signal part, calledlookahead, and not on the AMR-WB decoder.

In fact, in the AMR-WB coder, each new 20 ms input frame at 16 kHzcorresponds to a time segment corresponding to the last 15 ms of thecurrent frame to be coded by ACELP model and 5 ms of future signal (orlookahead). The first 5 ms of the current frame to be coded have alreadybeen received and stored as “lookahead” of the preceding segment. Thelast 12 samples obtained after resampling from 16 to 12.8 kHz on thecoder therefore correspond to the last samples of the 5 ms future signalat 12.8 kHz. Consequently, the current 20 ms frame at 12.8 kHz (i.e. 256samples) and the 5 ms of future signal (i.e. 64 samples) is complementedwith 5 ms of past original signal (loopback) to form the LPC analysisbuffer of 384 samples (30 ms) which is weighted by an LPC analysiswindow of the same length.

The last 12 samples of the “lookahead” at 12.8 kHz comprising aresampling error have a very low relative weight in the window used forthe linear prediction (LPC), and a fortiori they have impact only on theestimated LPC envelope and this impact is very negligible. It isimportant to note that the 12 erroneous samples are replaced by the“exact” resampling values on the next frame, the error is thereforepresent only temporarily in the current frame for the future signal(lookahead) and affects only the LPC analysis. Thus, the delaycompensation technique of the AMR-WB coder does not affect the coding ofthe waveform of the signal in the current frame in the AMR-WB codec.This mode will hereinafter be referred to as: “use by frame with futuresignal”. The samples that are thus generated are only used temporarilyfor intermediate calculations (LPC analysis) and are replaced by thesamples correctly resampled when the signal of the next frame is known.It will be noted that, in this configuration, for an output frame oflength lg_out for each frame, lg_out+12 samples are generated by theresampling.

This delay compensation technique used on the AMR-WB coder is notapplied to the AMR-WB decoder.

Thus, the codec (coder+decoder) has a total algorithmic delay of 25.9375ms due to the coder (20 ms frame+5 ms lookahead) and to the resamplingon the decoder (0.9375 ms).

The delay compensation technique of the AMR-WB coder could not be usedto reduce the QMF filtering delay of the G.722 codec, because it wouldgreatly degrade the quality of the coding signal. In effect, in theG.722 codec, the samples resulting from the filtering (the low-band andhigh-band signals) directly constitute the input signals of the twoADPCM sub-codecs which operate without “lookahead” and which do not makeit possible to correct these values from one frame to another. This modewill hereinafter be referred to as: “continuous frame-by-frame use”.

SUMMARY

An exemplary embodiment of the present application relates to a methodfor resampling an audio frequency signal in an audio frequency signalcoding or decoding. The method is such that it comprises the followingsteps for each signal block to be resampled:

-   -   determinating, by adaptive linear prediction, a number of future        signal samples, this number being defined as a function of a        chosen resampling delay;    -   constructing a resampling support vector from at least samples        of the current block and determined future signal samples;    -   applying a resampling filter to the samples of the resampling        support vector.

Thus, the resampling according to the invention makes it possible toreduce the resampling delay by filter (per block or per frame), whilekeeping a performance level very close to a continuous resampling. Theprediction of the future signal in each block or frame makes it possibleto have samples closer to the real samples than if these future sampleswere set at a null value. The performance of the resampling process forthese latter samples is therefore better.

This resampling method is also adaptive since it is possible to adaptthe number of future signal samples to be taken into account as afunction of the desired delay. For a resampling without delay, thenumber of future signal samples to be determined then corresponds to thedelay of the resampling filter. If a delay lower than the delay of thefilter is tolerated, then the number of future signal samples cancorrespond to just a part of the delay of the resampling filter.

With the adaptation of the resampling delay being performed per signalblock, it is then possible to easily switch, from one block to anotheror from one frame to another, between different resamplingconfigurations (including the FIR filter used for this purpose) or toswitch from a direct coding of a given frame at a certain samplingfrequency to a coding of the next frame with resampling, or vice versa.

The various particular embodiments mentioned hereinbelow can be addedindependently or in combination with one another, to the steps of theresampling method defined above.

In a simple embodiment, the step of determination by adaptive linearprediction comprises the following steps:

-   -   obtaining coefficients of a linear prediction filter of        predetermined order;    -   obtaining future signal samples by application of the prediction        filter obtained to an excitation signal of null value.

In a particular embodiment, the coefficients of the linear predictionfilter are obtained by reading parameters stored in the coding ordecoding step.

Thus, when the coding module, independently of the resampling device,comprises an LPC analysis which already determines the parameters of theprediction filter, there is no need to recalculate these parameters inthe resampling method. It is sufficient to merely read the parameterswhich have been stored (quantified or not).

In another embodiment, the coefficients of the linear prediction filterare obtained by analysis from at least samples of the current block.

The LPC analysis is then done directly in the resampling device.

In an exemplary embodiment, the linear prediction is performed on anaudio frequency signal on which a pre-emphasis processing has beenperformed.

The pre-emphasis makes it possible to ensure a better digital stabilityin a fixed-point notation implementation, in particular for the signalshaving a strong slope and spectral dynamic. It reduces the spectraldynamic of the signal, the distribution of the power of the signal overthe frequency bands thus becomes more uniform after the pre-emphasis.The post-pre-emphasis modeling parameters have a lower dynamic, and itis easier to ensure the stability of the system and also easier toimplement the algorithm using this model with a fixed-point notationarithmetic.

In possible embodiments, the adaptive linear prediction is a predictionfrom one of the following methods:

-   -   short-term linear prediction;    -   long-term linear prediction;    -   combination of short-term linear prediction and long-term linear        prediction;    -   erased frame concealment process.

Thus, any more or less accurate prediction type is possible for theimplementation of the method while ensuring an effectiveness in terms ofadvantageous signal-to-noise ratio.

The present invention also targets a device for resampling an audiofrequency signal in an audio frequency signal coder or decoder. Thedevice is such that it comprises:

-   -   an adaptive linear prediction module suitable for determining,        for each signal block, a number of future signal samples defined        as a function of a chosen resampling delay;    -   a module for constructing a resampling support vector from at        least samples of the current block and determined future signal        samples;    -   a resampling filter applied to the samples of the resampling        support vector.

This device offers the same advantages as the method describedpreviously, that it implements.

In a particular embodiment, the adaptive linear prediction modulecooperates with a prediction analysis module included in the predictioncoding or decoding module of the coder or decoder.

Thus, the complexity of the resampling device is reduced since there isno need to include any LPC analysis module. The parameters obtained fromthe analysis module of the coding or decoding module are stored in thecoding or the decoding and can thus be used in the resampling.

The present invention also targets an audio frequency signal coder anddecoder comprising at least one resampling device as described.

In a particular embodiment, the coder or the decoder comprises aresampling device using at least two resampling filters having differentdelays, at least one of the filters being implemented according to themethod as described previously for which the determination of the numberof future signal samples is a function of the delay difference of thetwo resampling filters used.

Since the resampling filters are often associated with different codingmodes, this embodiment makes it possible to switch easily from onecoding mode to another without there being any audible artifacts.

The invention targets a computer program comprising code instructionsfor implementing the steps of the resampling method as described, whenthese instructions are executed by a processor.

Finally, the invention relates to a processor-readable storage medium,incorporated or not in the resampling device, possibly removable,storing a computer program implementing a resampling method as describedpreviously.

BRIEF DESCRIPTION OF THE DRAWINGS

Other features and advantages of the invention will become more clearlyapparent from reading the following description, given purely as anonlimiting example, and with reference to the attached drawings, inwhich:

FIG. 1 illustrates the impulse response of a resampling filter in aprior art method as described previously;

FIG. 2 illustrates the impulse response of a resampling filter in aprior art method with compensation of the delay by zeros as describedpreviously;

FIG. 3 illustrates an exemplary audio signal coder comprising aresampling device according to an embodiment of the invention;

FIG. 4a illustrates, in flow diagram form, the steps of a resamplingmethod according to an embodiment of the invention;

FIG. 4b illustrates, in flow diagram form, the steps of a variantembodiment of a resampling method according to the invention;

FIG. 5a illustrates, in flow diagram form, the details of the step ofdetermination of the pre-emphasis factor according to an embodiment ofthe invention;

FIG. 5b illustrates, in flow diagram form, the details of the linearprediction step according to an embodiment of the invention;

FIG. 6 illustrates the form of an analysis window used in an embodimentof the invention;

FIGS. 7a to 7l show, for different sample positions following thecurrent signal frame to be resampled, a comparison of thesignal-to-noise ratio as a function of the center frequency obtained bythe application to a test signal of the resampling method of the priorart implemented in the AMR-WB coder and of the resampling methodaccording to a first and a second embodiment of the invention;

FIGS. 8a to 8c show, for different sample positions following thecurrent signal frame to be resampled, a comparison of thesignal-to-noise ratio for three different cases of change of resamplingfrequencies according to the resampling method of the prior artimplemented in the AMR-WB coder and of the resampling method accordingto a first and a second embodiment of the invention; and

FIG. 9 illustrates an example of an audio signal decoder comprising aresampling device according to an embodiment of the invention;

FIG. 10 illustrates a physical representation of a resampling deviceaccording to an embodiment of the invention.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

FIG. 3 illustrates an example of an audio coder comprising a resamplingdevice 300 according to an embodiment of the invention.

The codec illustrated here is a coder of audio signals (mono), multi-bitrate (with bit rates set from 7.2 to 128 kbit/s) operating at the inputand output sampling frequencies of 8, 16, 32 or 48 kHz. Interest isfocused first of all on the coder part represented in FIG. 3, theassociated decoder being described later in conjunction with FIG. 9.

The input signal is divided into 20 ms frames (block 310), each framecan be coded either according to a time domain (TD) approach of CELPtype (309) or according to a frequency domain (FD) approach of MDCT type(block 312) before being multiplexed by the multiplexing module 313. Itis considered here that the codings of CELP and MDCT type are known tothose skilled in the art. The choice of the mode (block 311)—whichincludes that of the internal coding frequency—is not detailed here.

In this codec, which is flexible in terms of bit rates and of samplingfrequencies, a number of resampling configurations of a frequency“f_(In)” (In for input) at a frequency “f_(Out)” (Out for output) arenecessary. In an embodiment described here, the configurations used arelisted in table 1 below:

TABLE 1 Config. Conversion configuration filt_(—) fac_(—) fac_(—) number(f_(In) −> f_(Out)) FIR filter used len_(in) num den 1   8000 Hz −> 12800 Hz f_5_8_129 16 8 5 2 12 800 Hz −> 8000 Hz f_5_8_129 24 5 8 3 16 000Hz −> 8000 Hz f_12_180 30 6 12 4 12 800 Hz −> 16 000 Hz f_15_180 12 1512 5 12 800 Hz −> 32 000 Hz f_15_180 12 15 6 6 12 800 Hz −> 48 000 Hzf_15_180 12 15 4 7 16 000 Hz −> 12 800 Hz f_15_180 15 12 15 8 16 000 Hz−> 32 000 Hz f_12_180 15 12 6 9 16 000 Hz −> 48 000 Hz f_12_180 15 12 410 32 000 Hz −> 12 800 Hz f_15_180 30 6 15 11 32 000 Hz −> 16 000 Hzf_12_180 30 6 12 12 48 000 Hz −> 12 800 Hz f_15_180 45 4 15 13 48 000 Hz−> 16 000 Hz f_12_180 45 4 12In this table, the frequency values in bold indicate “external”frequencies (that is to say input and/or output frequencies) of thecodec, and the other frequency values are “internal” samplingfrequencies for the coding of the low band of the signal—in a mannersimilar to the AMR-WB coding which has only one external frequency at 16kHz and one internal frequency at 12.8 kHz. “filt_len_(in)” representsthe length of the filter, “fac_num” represents the up-sampling factorand “fac_den” the down-sampling factor.

With no loss of generality, the FIR filters are designed in the exampleillustrated here according to the conventional method, called “windowmethod”, because it involves a windowing of a cardinal sine (sin(x)/x).The FIR filters are for example designed as explained below.

For example, the filter f_5_8_129 is obtained with the following matlabcommand:

f_5_8_129=[0 0 0 0 fir1(248,(3775/32000),hanning(249))*4.999895 0 0 00];

with a cutoff frequency (−6 dB) at 64 kHz of 3775 Hz.

These coefficients are used as a filter of 16 coefficients at 8000 Hz(i.e. 128 coefficient at 64 000 Hz) and as a filter of 24 coefficient at12 800 Hz (i.e. 120 coefficients at 64 000 Hz, disregarding the lastvalues)

The filter f_12_180 is obtained with the following matlab commands:

ftmp=fir1(358,1/12,hanning(359));

f_12_180=[0 ftmp/ftmp(180) 0];

with a cutoff frequency (−6 dB) at 192 kHz of 8000 Hz.

The filter f_15_180 is obtained with the following matlab commands:

ftmp=fir1(358,1/15,hanning(359));

f_12_180=[0 ftmp/ftmp(180) 0];

with a cutoff frequency (−6 dB) at 192 kHz of 6400 Hz.

In variants of the invention, of course, other FIR filter design methodscan be used.

These conversion configurations are justified here, with no loss ofgenerality, by the use of 2 internal frequencies, 12.8 kHz and 16 kHz,in the coding algorithm. Neither the way in which the choice of theinternal sampling frequency (12.8 or 16 kHz) is made, nor the way inwhich the choice of the type of coding to be employed (block 311) ismade is detailed here. That exceeds the scope of the invention. However,it will be remembered that the choice of the internal frequency can bemade independently in each frame, for a same input and/or outputfrequency of the codec, which, for example, means that it is possible,at the frame N to use a resampling according to a configuration i, atthe frame N+1, a resampling according to the configuration j differentfrom i (but with a same “external” frequency), and at the frame N+2, noresampling, which means a direct coding of the frame at the inputfrequency f_(In)—in practice, this last case is possible in thepreferred embodiment only in the following situations:

if the coding mode chosen is the coding of FD type, which alwaysoperates at the frequency f_(In)

if the coding mode chosen is the TD coding and the input frequencyf_(In) corresponds to the internal TD coding frequency.

However, in a variant of the invention, the coding of FD type will beable to be forced to operate at the same internal coding frequency asthe TD coding, in order to facilitate the switching thereof.

Note that the transition from a TD coding to an FD coding and vice versais not described here because it goes beyond the scope of the invention.

When the sampling frequency of the input signal f_(In) is greater thanthe internal coding sampling frequency the coding algorithm TD COD or FCCOD provides the coding of the signal in high band (frequencies greaterthan 6.4 or 7 kHz), this coding of the high band is not detailed here.

Thus, the coder comprises a resampling device 300 which includes aparameterizable resampling filter because it can operate with a numberof FIR filter coefficient configurations (block 305). In the embodimentsdescribed hereinbelow, the resampling filter is a polyphase filter. Theinvention applies also to other types of implementation of resampling byFIR filter such as, for example, a resampling filter of non-optimalcomplexity which does not involve the polyphase representation.Moreover, the invention applies also for other sampling frequencyconversion ratios.

With the exception of the first three configurations (numbers 1 to 3),all the other configurations use a polyphase filter of FIR type with adelay of 0.9375 ms (12 samples at 12.8 kHz, 15 samples at 16 kHz, 30samples at 32 kHz and 45 samples at 48 kHz).

The polyphase resampling filtering (block 305) in configurations 4 to 13is performed according to an algorithm derived from the polyphaseresampling by FIR defined in the ITU-T G.718 codec (see the realizationin the source code of G.718 in the “modify_fs.c” file).

The interest here is focused on the first 3 configurations involving an“external” sampling frequency of 8000 Hz. For these configurations, alonger FIR filter is necessary to have adequate filtering performance,in particular to guarantee a sufficient rejection of spectral images orof the spectral aliasing which can occur in the frequencies where theear is very sensitive.

Without the implementation of the resampling method of the invention,these 3 configurations would result normally in 25 samples of delay at12.8 kHz for the case of the resampling from 8000 Hz to 12 800 Hz, 15samples of delay at 8 kHz for the cases of the resampling from 12 800 Hzto 8000 Hz and of the resampling from 16 000 Hz to 8000 Hz. Generally,the delay at the output sampling frequency is rounded to the integerbelow filt_len*fac_num/fac_den, where filt_len is the length of thefilter, fac_num is the up-sampling factor and fac_den is thedown-sampling factor (see also in table 1), but it would also bepossible to consider a delay with a fraction of half a sample.

The implementation, in the resampling device 300, of the resamplingmethod according to the invention and described below with reference toFIGS. 4a and 4b , makes it possible, in the case of this coder:

-   -   to limit the effective delay of configurations 1 to 3 to obtain        a delay identical to the other conversion configurations 4 to 13        (which have a delay of 0.9375 ms). To do this, the resampling        device comprises an adaptive linear prediction module 301,        suitable for determining, for each signal frame, a number of        future signal samples defined as a function of a chosen        resampling delay. It will be noted that the number of samples is        theoretically parameterizable but in practice, it is kept        constant for the defined codec configuration.    -   To be able to switch all the defined resampling configurations,        even if the associated theoretical delay is different.

In a particular embodiment of the invention which is not described intable 1, the coder can comprise a number (at least two) of polyphase FIRresampling filters resulting in different delays. For example, in amultiple bit rate coder, for an input which is always at 16 000 Hz, itis possible to use 2 different internal frequency coding cores accordingto the bit rate: 8000 Hz for the lower bit rates and 12 800 Hz for thehigher bit rates. To have a sufficient filtering performance level, inparticular to guarantee a sufficient rejection of spectral images or ofthe spectral aliasing, the resampling from 16 000 Hz to 8000 Hz requiresa longer FIR filter than the resampling from 16 000 Hz to 12 800 Hz.These two filterings therefore have a different delay. To be able toswitch between these two coding modes without artifact (in case ofchange of bit rate), these delays must be harmonized (made equal). Ifthe length of the resampling FIR filter is reduced from 16 000 Hz to8000 Hz, the quality is generally degraded because the spectral aliasingwould not be sufficiently well attenuated and would become audible. Ifthe length of the resampling FIR filter is increased from 16 000 Hz to12 800 Hz, or an additional delay is applied to the resampled signal,the overall delay of the coding/decoding is increased, which can hamperthe interactivity.

By using the resampling method of the present invention, it is possibleto reduce the delay of the longer FIR filterings to the level of theshorter filtering delay, without notable loss of quality, compared withthe original filtering. In fact, the simulation results show that thesignal-to-noise ratios are very high between the normal filtering andthe low-delay filtering according to the present invention. It is alsodemonstrated by the listening tests that the difference between thesignals obtained with the normal filtering and the low-delay filteringaccording to the present invention is not audible.

Thus, in the case presented here, the coder (or even the decoder)comprises two polyphase resampling filters with different delays. Atleast one of the resampling filters is a filter implemented as describedpreviously according to the invention in which the determination of thenumber of future signal samples is a function of the delay difference ofthe two resampling filters used in the two devices.

For example, in table 1, the configurations 3 to 7 can be used for anexternal frequency of 16 000 Hz and internal frequencies of 8000 Hz and12 800 Hz. In this case, it can be seen that the delay at the inputfrequency (filt_len) is respectively 30 and 15 samples; it is thereforenecessary to predict the difference, i.e. 15 samples at 16 000 Hz, to beable to reduce the delay of the configuration 3 to the level of thedelay of the configuration 7. The invention detailed later will be ableto be used to reduce the delay of the configuration 3 and to be able toalternate between the configurations 3 and 7 transparently, because theythen have the same delay of 15 samples.

To revert to the example of the coder of FIG. 3 and of theconfigurations listed in table 1:

in the case of the resampling from 8000 Hz to 12 800 Hz, it is necessaryto reduce the delay from 25 to 12 samples, i.e. generateplus_sample_out=13 additional samples at 12 800 Hz, which necessitatesthe extrapolation of plus_sample_in=8 samples at 8000 Hz.

For the case of the resampling from 12 800 Hz to 8000 Hz, it isnecessary to reduce the delay from 15 to 7 samples, i.e. generateplus_sample_out=8 additional samples at 8000 Hz, which necessitates theextrapolation of plus_sample_in=12 samples at 12 800 Hz.

For the case of the resampling from 16 000 Hz to 8000 Hz, it isnecessary to reduce the delay from 15 to 7 samples, i.e. generateplus_sample_out=8 additional samples at 8000 Hz, which necessitates theextrapolation of plus_sample_in=15 samples at 16 000 Hz. It should benoted that, at 8000 Hz, 0.9375 ms corresponds to 7.5 samples that havebeen rounded down to 7 samples.

The resampling device 300 illustrated in FIG. 3, receives audio signalblocks as input, and in this embodiment they are 20 ms frames receivedby the block 310 which also has in memory a set of samples from pastframes.

This resampling device comprises an adaptive linear prediction module301 suitable for determining, for each signal block or frame, a numberof future signal samples defined as a function of a chosen resamplingdelay.

This predicted number of future signal samples is used to determine theresampling support defined by the module 304 for constructing theresampling support vector. This resampling support vector is, forexample, a concatenation of possible past signal samples, samples fromthe current block or frame and future signal samples predicted by themodule 301. The past signal samples serve as memory for the resamplingFIR filter.

Nevertheless, the construction of this support vector also comprises thefollowing implementation:

-   -   the past signal can be stored in the memories of the resampling        FIR filter and is not therefore directly concatenated with the        samples of the current frame (but the signal of the current        frame is indeed the continuity of these memories containing the        past signal)    -   the predicted future signal can also be stored in a separate        vector and its resampling can be done separately from that of        the signal of the current frame, as long as the necessary        memories are updated correctly. In the case of the separate        resampling of the future signal, the memories of the resampling        filter are initialized by the latest samples of the current        frame. Once again, despite this separation, the predicted future        signal is indeed the continuation of the signal of the current        frame.

In this document, with no loss of generality, the term “construction ofthe support vector” also covers the cases where the signals are notreally copied one after the other in a same vector but stored indifferent vectors.

The filter 305 is then applied to this resampling support vector toobtain a signal resampled at the desired output frequency.

The linear prediction module 301 can comprise a short-term predictionanalysis module 302 (LPC) suitable for determining the coefficients of alinear prediction filter as described later in relation to FIG. 4a .This LPC analysis module (302 b) can, in another advantageousembodiment, be included in the prediction coding module 309 of thetemporal coding of TD type (for example a CELP coding). Thus, the sameanalysis module can be used both to predict useful future samples forthe resampling device and to code the signal before transmission. Thistherefore reduces the complexity of the resampling device whichcooperates with the analysis module of the coding module.

The module 301 further comprises a prediction filtering module 303 bythe 1/A(z) filtering of a null signal to obtain a set of future samplesbuf_(fut).

FIG. 4a therefore illustrates the main steps of a resampling methodaccording to an embodiment of the invention.

The steps of this method are implemented with, as input (Buf_(In)),frames of lg samples at the input sampling frequency f_(In). There isalso access to the past samples of this input signal through thememories. From this input signal, the step E401 determines the number ofsamples to be predicted plus_sample_in, as a function of the desireddelay and predicts this number of future signal samples by linearprediction. The result of this prediction is concatenated on the inputsignal (current frame and past frames for the memories) in the step E402of construction of the resampling support vector. This support vector istherefore, in one embodiment, a concatenation of samples of the pastsignal, samples of the current frame and determined future signalsamples.

In the step E403, the resampling filtering is performed by applicationof a resampling filter, for example with finite impulse response (FIR),to the samples of the resampling support vector, and the resampledsignal buf_(out) is supplied as output, at the output resamplingfrequency f_(out).

A number of cases are then possible:

-   -   in the case of a continuous frame-by-frame use (lg samples as        input), only the last lg_out=lg*fac_num/fac_den samples are        calculated.    -   In the case where more samples have to be obtained (use per        frame with future signal) (lg+plus_sample_in samples as input),        as is the case in the coder of the AMR-WB standard,        lg_out+plus_sample_out samples are calculated, where        plus_sample_out=plus_sample_in*fac_num/fac_den. In fact, in the        AMR-WB coder, the current 20 ms frame is resampled and 15        additional samples are resampled; the first step replaces the        error from resampling performed in the second step. In this        example, lg=320, plus_sample_in=15.    -   The use can also be one-off, when, for example, a memory (a        piece or block of the signal) is resampled, for example in the        case of a switch between two coding modes. In this case, the        resampling input is not a frame (for example of 20 ms) but a        signal block. In order to apply the invention, it is important        to note that it is essential to have either the past of the        block to be converted or else an LPC model already        pre-calculated from the past—it will be noted that, with a        coding of TD type already using a linear prediction, it is        generally possible to store the parameters (LPC or equivalent        coefficients) calculated and/or coded in the TD coder and/or        decoder in each frame. Thus, in a variant of the invention,        these LPC parameters will be able to be reused, which simplifies        the implementation of the block 302 since it then involves a        simple looking up of stored values (possibly quantified).        In parallel, the memory of the resampling filter is updated in        E405. Once again, a number of cases are possible:    -   in the case of a continuous use with more samples to be        generated, as is the case in the AMR-WB standard, the last        mem_len samples of the input frame are stored, without the        predicted samples: mem_sig(0 . . . mem_len−1)=frame(lg−mem_len .        . . lg−1), upon resumption of the resampling, the samples        obtained at the output sampling frequency replace the samples        obtained by using the predicted input signal.    -   In the case of continuous frame-by-frame use, the samples        obtained by using the predicted input signal are not replaced,        only lg_out samples are calculated at the output sampling        frequency. If it is considered that the new frame begins at the        sample of index lg+plus_sample_in, the memory of the resampling        FIR filtering is made up of the past samples of index        (lg+plus_sample_in−mem_len lg+plus_sample_in−1) of which a part        of this memory, of index (lg . . . lg+plus_sample_in−1), can be        either the true signal or the predicted signal. By using the        true signal, the first samples are equal to those obtained with        a filtering without prediction (result considered as optimal)        but, between the last sample obtained with the prediction during        the preceding frame and the first sample obtained with the true        signal, it is possible to have a small discontinuity. In case of        use of the predicted signal in the memory, there is no        discontinuity, but a slight error spreads over another filt_len        samples. In the preferred embodiment, the first solution is used        because this slight discontinuity is not audible.    -   In case of one-off use, the memory update is not necessary after        the resampling, but the resampling memories must be initialized        before the resampling operation, with the corresponding past        input signal.

In a variant, the LPC analysis used to predict the future signal isperformed not on the signal directly in the current frame, but on thepre-emphasized signal, obtained from the filtering of the current frameby a filter of the form 1-μ·z⁻¹, where μ is calculated adaptively or setat a predetermined value. This variant is illustrated in FIG. 4b .Compared to FIG. 4a , a step E406 of determination of the pre-emphasisfactor μ is added. By using this factor μ, the input signal ispre-emphasized in this step E407 by 1-μ·z⁻¹ filtering. It should benoted that this filtering necessitates a memory sample, therefore, inthis variant, the size of the memory is to be increased by 1. Thepre-emphasized signal is the input of the steps E401 and E402. Theconcatenated signal is then de-emphasized by using the same factor μ inthe step E408 by 1/(1−μz⁻¹) filtering. It should be noted that, for agiven signal, the sequencing of the pre-emphasis before LPC analysisfollowed by a de-emphasis by the same factor μ is transparent, that isto say that precisely the input signal is retrieved. Therefore, if thesignal is stored before the pre-emphasis, only the predicted part has tobe de-emphasized to reduce the calculation complexity. Thisde-emphasized predicted part is then concatenated on the stored signalto form the resampling support vector.

There are a number of techniques for determining the pre-emphasis factorμ for which values are between −1 and 1.

-   -   μ can be constant, for example μ=0.68    -   μ can be constant, dependent on the input sampling frequency    -   μ can be adaptive according to an analysis of the tilt of the        spectrum (method known from the prior art).

FIG. 5a illustrates this step E406 of FIG. 4b , of determination of thepre-emphasis factor. In the step E501, the signal is windowed by ananalysis window. In the step E502, a self-correlation of order M=1 (r(0)and r(1)) is calculated and a noise threshold (or noise floor) isapplied to r(0) in the step E503, to avoid the arithmetical problems ofthe low level input signals.

These steps of self-correlation, of application of a noise threshold,are for example described in ITU-T recommendation G.729 subsection3.2.1.

The calculations of self-correlations r(k) with a window of length L forthe shifts k=0, . . . , M are of the form:

${{r(k)} = {\sum\limits_{n = k}^{L - 1}\;{{s_{w}(n)}{s_{w}( {n - k} )}}}},{k = 0},\ldots,M$in which s_(w)(n)=s(n)·w(n) and s(n) corresponds to the last L samplesof the signal of the current frame and possibly of past signal if thelength L is greater than the length of the current frame.

In the preferred embodiment an LPC window w(n) length L=240 is used, anexample of which is illustrated in FIG. 6.

It can be seen that the form of this window is asymmetrical with theweight concentrated on the end of its support (on the most recentsamples). The matlab commands to construct this window with L=240 arefor example given below:

L1 = L−8; for i = 0: (L1−1) w(i+1) = 0.54 − 0.46 * cos(2 * i * pi / (2 *L1 − 1)); end for i = L1 : (L−1) w(i+1) = cos((i − L1) * 2 * pi / (31));end

In variants of the invention, other values of the LPC order M, otherforms and lengths of LPC window will be able to be used without changingthe nature of the invention. The “noise floor” will be able to be usedin a conventional manner by multiplying the first correlationcoefficient by a factor >1 or by limiting the value of this firstcoefficient to a minimum value.

Finally, the factor is calculated in the step E504 as μ=r(1)/r(0).

FIG. 5b describes, in more detail, an embodiment of the step E401 of theFIG. 4, of linear prediction to determine the future samples accordingto the invention.

For example, this step E401 can comprise a step E506 of calculation ofcoefficients of a linear prediction filter of predetermined order, fromthe samples of the current frame and possibly samples of the precedingframes and a step E507 of obtaining future signal samples by applicationof the calculated prediction filter to an excitation signal of nullvalue.

The steps E501, E502 and E503 of FIGS. 5a and 5b are similar, but with adifferent prediction order M. The other parameters such as the form orthe length of the analysis window or even the “noise floor” can also bedifferent. In both cases, the modules in common can be used to reducethe complexity.

More specifically, the input signal (pre-emphasized or not) is windowedin the step E501. It is for example possible to use the same type ofwindow as that illustrated in FIG. 6. The self-correlation function iscalculated at the chosen order (in the example, M=10) in E502 and anoise floor is applied to r(0) in the step E503, as described forexample in subsection 3.2 of the G.729 standard.

In the step E505, a step called “Lag windowing” (a method known to thoseskilled in the art) is performed, also described notably in thesubsection 3.2.1 of the G.729 standard.

This step of “Lag windowing” for the input sampling frequency (f₁) is ofthe form:r(i)=r(i)*w _(lag)(i), i=0, . . . ,M

in which the coefficients w_(lag) (i) are defined as follows:

${{w_{lag}(i)} = {\exp\lbrack {{- \frac{1}{2}}( \frac{2\pi\; f_{0}i}{f_{s}} )^{2}} \rbrack}},{i = 1},\ldots,16$in which f_(s)=f_(In) is the frequency of the signal to be resampled andin which, for example, f₀=60 Hz.

In the step E506 (implemented by the module 302 of FIG. 3), thecoefficients A[i], i=0, . . . , M, of a linear prediction filter A(z) oforder M, are calculated by the Levinson-Durbin algorithm as describedwith reference to subsection 3.2.2 of G.729 or subsection 6.4.3 of theAMR-WB standard. In the preferred embodiment, an LPC order M=10 is used.

In the step E507 (implemented by the module 303 of FIG. 3), thesynthesis filter 1/A(z) is applied to a null signal to give a predictionof the future signal samples. This prediction is performed recursively,by 1/A(z) filtering with null input (filter excitation signal), forplus_sample_in samples at the end of the frame of length lg (i=lglg+plus_sample_in −1):

${{sig}(i)} = {\sum\limits_{j = 1}^{M}\;{{- {{sig}( {i - j} )}}*{\alpha(j)}}}$

In a variant of the invention, other methods for calculating linearprediction coefficients will be able to be used, for example it will bepossible to use the Burg method implemented, for example, in the SILKcoder known from the prior art.

In another variant, the linear prediction coefficients will be able tobe estimated by an approach of LMS (Least Mean Squares) or RLS(Recursive Least Squares) type of adaptive filtering.

In another alternative, the LPC coefficients will be able to be directlyobtained from an analysis and/or quantification of the associatedparameters, performed on the signal in the coder of TD type (309) usingan LPC prediction (302 b) even in the FD coder, provided that a linearprediction is performed in the FD coder.

For example, in the CELP decoder of the AMR-WB codec there are LPCcoefficients (of order 16) in each subframe and it is in particularpossible to use LPC coefficients decoded in the last subframe to predictthe future decoded signal and thus eliminate the delay of the resamplingof the CELP decoder.

In another variant, the null excitation (null input) in the step E507can be replaced by an excitation predicted, for example, by pitchprediction in the excitation domain.

In other variants of the invention, the (short-term) linear predictionwill be replaced by (long-term) pitch prediction in the domain of thesignal, this prediction may be fractional or multi-tap.

It will be noted that it would be possible to perform the prediction inthe frequency domain instead of a temporal approach; however, thisalternative approach in the frequency domain requires an analysistransformation (for example FFT), a prediction of the future spectrum,for example by repetition of the amplitudes and continuity of the phasesof the most important spectral rays and an inverse synthesistransformation or a sinusoidal synthesis; this alternative is generallymore complex than the temporal approach described previously, all themore so as the frequency analysis has to have a temporal support that islong enough to have a frequency resolution sufficient to identifyspectral rays (tones). This approach is not ideal when the aim is toextrapolate a limited number of samples (less than the frame length).

In yet another embodiment, the adaptive linear prediction describedpreviously can be replaced by an erased frame concealment process inorder to extrapolate the future signal by a more sophisticated signalmodel. Such a technique is for example described in the European patentpublished under the number: EP1 316 087.

In other variants of the invention, the resampling by FIR filter will beable to be replaced by other resampling methods by IIR filtering orpolynomial interpolation. In this case, the principle remains the same:the future signal is predicted and the resampling is applied by takinginto account the future signal. In one embodiment, the case of 2resampling configurations with different delays is considered and theinvention makes it possible to bring the longest delay to the lowestdelay value.

To be able to demonstrate the effectiveness of low-delay resamplingaccording to the method of the invention described previously in theexample of resampling from 8000 Hz to 12 800 Hz, a test signal is usedconsisting of a mix of 10 sinusoids, the frequency of which changes eachsecond. For the signal of the i^(th) second, the frequencies of thesesinusoids have been chosen randomly, around a center frequencyfe_(center)(i), in the interval [fe_(center)(i)−600, fe_(center)(i)+600]and fe_(center)(i)=500+100*i Hz, i=1 . . . 28.

FIGS. 7a to 7l represent the results of a comparison between theresampling method of the prior art in AMR-WB (dotted line), that of themethod according to the invention with a prediction filter of order M=4with analysis window of 20 samples (chain-dotted line) and that of themethod according to the invention with a linear prediction filter oforder 10 with analysis window of 240 samples (continuous line).

The figures represent the signal-to-noise ratio as a function of thecenter frequency of the test signal.

Each figure corresponds to a different position of the sample relativeto the end of the conventional frame obtained with a conventionalfiltering (which corresponds to the numbering #1, . . . , #12 of FIG.2). For example, FIG. 7a represents the signal-to-noise ratio (SNR) forthe samples in second position after the end of the conventional frame.FIG. 7b represents the signal-to-noise ratio for the predicted sample in3^(rd) position after the current frame, etc. FIG. 7l thereforerepresents the signal-to-noise ratio for the predicted sample in 13^(th)position after the current frame.

It can be observed that the SNR decreases with the increase in theposition because predicted samples are increasingly used during thefiltering and that for the same position, the SNR decreases with theincrease in the center frequency because the high frequencies are lesspredictable. However, in all cases, it is observed that the methodaccording to the invention, even with low prediction order, issignificantly more efficient than the method used in the AMR-WB coder.

The advantage of the use of a low order prediction is its low complexityand the ease of implementation of the calculations, above all infixed-point notation arithmetic. The higher the order, the more thecomplexity increases and, at the same time, the more difficult itbecomes to ensure the stability of the filter.

FIGS. 8a to 8c show the same type of result over a very wide speechsignal base. Therein the SNR is seen as a function of the position ofthe sample for 3 different cases: from 8000 Hz to 12 800 Hz in FIG. 8a ,from 12 800 Hz to 8000 Hz in FIG. 8b and from 16 000 Hz to 8000 Hz inFIG. 8c . Once again, the algorithm according to the invention issignificantly more efficient than that used in the prior art (AMR-WB),even with low prediction order with short window.

FIG. 9 illustrates an example of an audio decoder comprising aresampling device 300 according to the invention. The resampling deviceis the same as that described with reference to FIG. 3.

The decoder illustrated here is a decoder of (mono) audio signals,multiple bit rates (with bit rates set from 7.2 to 128 kbit/s) operatingat the output sampling frequencies of 8, 16, 32 or 48 kHz.

Based on the frame received and demultiplexed (block 901), the output isswitched (904) between the output of a time division decoder (TD DEC) ofCELP type (902) using a linear prediction (902 b) and a frequency domaindecoder (FD DEC).

FIG. 10 represents an example of hardware embodiment of a resamplingdevice 300 according to the invention. The latter can be made anintegral part of an audio frequency signal coder, decoder or of anequipment item receiving audio frequency signals.

This type of device comprises a processor PROC cooperating with a memoryblock BM comprising a storage and/or working memory MEM.

Such a device comprises an input module E suitable for receiving audiosignal frames Buf_(In) at a sampling frequency f_(In).

It comprises an output module S suitable for transmitting the resampledaudio frequency signal Buf_(out) at the sampling frequency of f_(Out).

The memory block can advantageously comprise a computer programcomprising code instructions for implementing the steps of theresampling method within the meaning of the invention, when theseinstructions are executed by the processor PROC, and in particular thesteps of determination by adaptive linear prediction of a number offuture signal samples, this number being defined as a function of achosen resampling delay, of construction of a resampling support vectorfrom at least samples of the current block and determined future signalsamples, of application of a resampling filter to the samples of theresampling support vector.

Typically, the description of FIG. 4a repeats the steps of an algorithmof such a computer program. The computer program can also be stored on amemory medium that can be read by a reader of the device or that can bedownloaded into the memory space thereof.

The memory MEM stores, generally, all the data necessary to implementthe method.

Although the present disclosure has been described with reference to oneor more examples, workers skilled in the art will recognize that changesmay be made in form and detail without departing from the scope of thedisclosure and/or the appended claims.

The invention claimed is:
 1. A method comprising acts of: resampling anaudio frequency signal by an audio frequency signal coding or decodingdevice, wherein resampling comprises the following acts for each signalblock to be resampled: determining, by adaptive linear prediction, anumber of future signal samples, the number being defined as a functionof a chosen resampling delay; constructing a resampling support vectorfrom at least samples of a current signal block and determined futuresignal samples; applying a resampling filter to the samples of theresampling support vector to form a resampled audio frequency signal;and outputting the resampled audio frequency signal for at least one oftransmission and storage in a memory.
 2. The method as claimed in claim1, wherein the determining act comprises acts of: obtaining coefficientsof a linear prediction filter of predetermined order; obtaining futuresignal samples by application of the linear prediction filter to anexcitation signal of null value.
 3. The method as claimed in claim 2,wherein the act of obtaining the coefficients of the linear predictionfilter includes an act of reading parameters stored by the coding ordecoding device during a coding or decoding of the audio frequencysignal.
 4. The method as claimed in claim 2, wherein the act ofobtaining the coefficients of the linear prediction filter includes anact analyzing at least samples of the current signal block.
 5. Themethod as claimed in claim 1, further comprising an act of pre-emphasisprocessing the audio frequency signal to obtain a processed audiofrequency signal, wherein the adaptive linear prediction is performed onthe processed signal audio frequency signal.
 6. The method as claimed inclaim 1, wherein the adaptive linear prediction is a prediction methodselected from one of the following methods: short-term linearprediction; long-term linear prediction; combination of short-termlinear prediction and long-term linear prediction; erased frameconcealment process.
 7. A device for resampling an audio frequencysignal in an audio frequency signal coder or decoder, wherein the devicecomprises: an adaptive linear prediction module determining, for asignal block, a number of future signal samples defined as a function ofa chosen resampling delay; a module constructing a resampling supportvector from at least samples of the current signal block and determinedfuture signal samples; a resampling filter applied to the samples of theresampling support vector to form a resampled audio frequency signal;and a memory configured to store the resampled audio frequency signal.8. The device as claimed in claim 7, wherein the adaptive linearprediction module cooperates with a prediction analysis module includedin a prediction coding or decoding module of the coder or decoder. 9.The device as claimed in claim 7, further comprising a furtherresampling filter to include two resampling filters including theresampling filter and the further resampling filter, the two resamplingfilters having different delays, at least one of the two resamplingfilters being implemented such that the number of future signal samplesis determined as a function of a delay difference of the two resamplingfilters.
 10. The device as claimed in claim 7, wherein at least one ofthe coder and the decoder comprises a further resampling filter, theresampling filter and the further resampling filter having differentdelays, at least one of the resampling and the further resamplingfilters being implemented such that the number of future signal samplesis determined as a function of a delay difference of the resampling andthe further resampling filters.
 11. A non-transitory processor-readablestorage medium, on which is stored a computer program comprising codeinstructions for executing a resampling method, when the instructionsare executed by a processor of an audio frequency signal coding ordecoding device, wherein the code instructions configure the processorto perform acts of: resampling an audio frequency signal, comprising thefollowing acts for each signal block to be resampled: determining, byadaptive linear prediction, a number of future signal samples, thenumber being defined as a function of a chosen resampling delay;constructing a resampling support vector from at least samples of thecurrent signal block and determined future signal samples; and applyinga resampling filter to the samples of the resampling support vector toform a resampled audio frequency signal; and outputting the resampledaudio frequency signal for at least one of transmission and storage in amemory.